Transmit/receive beamforming signal generation

ABSTRACT

Transmit and/or receive beamforming signal generation includes a voltage-controlled oscillator (VCO) for generating a lower or higher master frequency output signal in accordance with a selection of a lower or higher frequency carrier frequency. A local oscillator generates local oscillator signals in quadrature in response to the maser frequency output signal. One or more mixer stages generate sidebands in response to a received information signal and the local oscillator signals in quadrature. The one or more mixer stages generate an output information signal in response to high-side injection of lower sidebands of the developed sidebands when the lower frequency carrier frequency is selected, and generate the output information signal in response to low-side injection of higher sidebands of the developed sidebands when the higher frequency carrier frequency is selected. Multi-band operation of transmit and receive arrays can be performed.

BACKGROUND OF THE INVENTION

Next-generation communication systems include multiple-input,multiple-output (MIMO) transceivers to accommodate high data throughputin wireless transmission systems. For example, transmitters in MIMOsystems are often arranged as an array of transmitter elements, whereeach transmitter element is coupled to respective antenna, and whereindividual antennas are often physically separated from an adjacentantenna by a fraction of the wavelength (e.g., a half wavelength) at aselected radio frequency (RF). Each of the transmitter elements in thearray can be phase shifted with respect to adjacent transmitter elements(e.g., contiguous transmitter elements arranged in a row or column). Thephase shifting of adjacent transmitter elements generates a highereffective output power based on constructive addition of electromagneticwaves in space (e.g., as compared to the sum of the average power out ofeach of the individual transmitter elements). Each of the transmitterelements is driven by a relatively complex RF circuit such that therelative cost of the RF circuits is a substantial portion of a systemcost.

SUMMARY

In described examples, a low power transmit and/or receive antenna arraysystem is arranged for enhancing effective power of a transmittedwaveform. In at least one embodiment, trigonometric weighted vectormodulation techniques for beam-focusing transmitter and phase-basedwaveform reconstruction are described. The techniques for trigonometricweighted vector modulation include (for example) baseband scalarweighting and quadrature-phased signals at LO (local oscillator) and RFfrequencies. Multi-band operation of the transmit and receive arrays areperformed in accordance with different sidebands at two or more LOfrequencies combined in response to quadrature signals generated bymoderate tuning range voltage-controlled oscillators (VCOs) such that,for example, a moderate tuning range VCO can be used in generatingsignals for a relatively wide range of operating frequencies. The localoscillator (LO) frequencies are generated coherently usingreconfigurable frequency divider and multipliers. The describedembodiment provides minimum signal loading at baseband frequencies,provides low distortion due to the current mode operation and sidebandcombination using up- and/or down-converter mixers, and permits lowerpower and area consumption for transmit/receive arrays.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1A is a schematic diagram of an architecture of a beamforming MIMOtransmitter 100.

FIG. 1B is a schematic diagram of various example embodiments of a phaseshifter 108 for generating LO signals and quadrature.

FIG. 2 is a schematic diagram of an architecture of a beamforming MIMOreceiver.

FIG. 3 is a schematic diagram of VCO polyphase quadrature signalgeneration circuits.

FIG. 4 is a schematic diagram of an architecture of a transmitterelements of a beam-focusing MIMO transmitter including cross-coupledquadrature mixer elements in accordance with example embodiments.

FIG. 5 is a schematic diagram of local oscillators (LO) for modulatingtransmitter elements of a beam-focusing MIMO transmitter in accordancewith example embodiments.

FIG. 6 is a schematic diagram of baseband phase shifter circuits inaccordance with example embodiments.

FIG. 7 is a schematic diagram of scalar elements for generatingtrigonometric weightage for a vector modulator in accordance withexample embodiments.

FIG. 8 is a schematic diagram of a one-local-oscillator up-conversionmixer in accordance with example embodiments.

FIG. 9 is a schematic diagram of a one-local-oscillator up-conversionmixer including programmable segmented transistors in accordance withexample embodiments.

FIG. 10 is a schematic diagram of a two-local-oscillator up-conversionmixer in accordance with example embodiments.

FIG. 11 is a schematic diagram of a transmitter elements forintermediate frequency-level trigonometric weighting in accordance withexample embodiments.

FIG. 12 is a schematic diagram of receiver elements for basebandtrigonometric weighting in accordance with example embodiments.

FIG. 13 is a schematic diagram of receiver elements for radio frequencytrigonometric weighting in accordance with example embodiments.

FIG. 14 is a flow diagram for selection of values for first and secondmultiplication-order operations for a local oscillator in accordancewith example embodiments.

DETAILED DESCRIPTION

Certain terms are used throughout the following description—andclaims—to refer to particular system components. Various names can beused to refer to a particular component (or system) wherein distinctionsare not necessarily made herein between components that differ in namebut not function. Further, a system can be a sub-system of yet anothersystem. In the following discussion and in the claims, the terms“including” and “comprising” are used in an open-ended fashion, andaccordingly are to be interpreted to mean “including, but not limited to. . . .” Also, the terms “coupled to” or “couples with” (and the like)are intended to describe either an indirect or direct electricalconnection. Thus, if a first device couples to a second device, thatconnection can be made through a direct electrical connection, orthrough an indirect electrical connection via other devices andconnections. The term “portion” can mean an entire portion or a portionthat is less than the entire portion. The term “components” can meanportions of a signal introduced into the signal in earlier (e.g.,“upstream”) operations upon the signal.

As described herein, MIMO- (multiple-input, multiple-output-) basedtransmitters and receivers achieve higher effective output and inputpower by controlling an array of multiple low power transmitter andreceiver elements for working concurrently and in parallel with likeelements. For example, transmitter elements include an “in-phase”digital-to-analog converter (DAC), an in-phase baseband analog filterfor modulating the output of the in-phase DAC, a quadrature DAC, aquadrature baseband analog filter for modulating the output of thequadrature DAC, an up-converting mixer for combining the outputs of thein-phase and quadrature baseband filters, and amplifiers for amplifyingthe output of the up-converting mixer and for coupling the amplifiedoutput to a respective antenna. Each of the number M of multiple lowpower transmitter elements works concurrently, for example, where eachpower transmitter element transmits a respective signal having adifferent phase shift from the signals being transmitted by adjacenttransmitter elements. (Adjacent transmitter elements can includecontiguous transmitter elements arranged in a row and/or column.) Thecooperation of the concurrently working multiple transmitter elementsincreases, for example, the effective transmitted power (e.g., viafocusing the transmitted beam) and/or the signal-to-noise ratio of thetransmitted waveforms.

Similarly, MIMO-based receivers include an array of receiver elements,where each power receiver element is coupled to a respective antenna inan antenna array, where individual antennas are physically separatedfrom an adjacent antenna by a faction of wavelength (e.g., halfwavelength). A receiver element includes, for example, amplifiers foramplifying the input signal received from a respective antenna, anin-phase filter for filtering the amplified input signal, an in-phaseanalog-to-digital converter (ADC) for digitizing the in-phase filteredamplified input signal and coupling the digitized in-phase values to adigital signal processor (DSP), a quadrature filter for filtering theamplified input signal, quadrature ADC for digitizing the quadraturefiltered amplified input signal and coupling the digitized quadraturevalues to the DSP. The DSP is a “back-end” processor arranged to phaseshift and combine each of the digitized quadrature values from eachrespective receiver element such that, for example, the effective gainof the antenna array is boosted.

For example, each of the M receiver elements in the receiver elementarray is selectively phase shifted for constructive combination byreceiver back-end (e.g., digital) processing of the signals received bythe selectively phase shifted M receiver elements in the receiverelement array. The constructive combination of the receiver back-endprocessing achieves a higher signal-to-noise (SNR) ratio for thecombined signal when compared to any one of the individually receivedsignals (e.g., based on each copy of the signals being correlated andconstructively added in-phase, while noise is uncorrelated). Spatialfiltering of the received signals is achieved, for example, byconstructive combination of a receiver back-end processing. Accordingly,the back-end processing results in higher effective signal-to-noiseratio as compared to the SNR of each of the individual waveforms of thereceived signal obtained from a respective receiver element.

MIMO transmitters and receivers (e.g., including transceivers) are usedextensively in emerging applications such as automotive radar as well asdata communication integrated circuits (ICs) for next-generationcommunication systems (e.g., 5G-level services). To improveefficiencies, communication systems using MIMO technologies aretypically implemented in a compact, power efficient manner. To improvecost effectiveness, a maximally reusable architecture is described asbeing scalable in accordance with certain designated multiple radiofrequency (RF) bands of usage (e.g., where emerging systems are designedfor operation within the 28 GHz and 35 GHz bands). Such systems areproposed to use array of between 16 and 512 antennas (e.g., inaccordance with a power of two) using large arrays of compact antennasfor handheld devices and base stations, where the handheld devices andthe base stations include tiles, where each tile contains four antennaarrays. In various embodiments, receivers and transmitters are describedfor processing two center frequencies (e.g., 28 GHz and 35 GHz).

As system complexity and levels of integration grow, MIMO-based systemsincreasingly employ transmitter and receivers of low power and low areafor achieving commercially viable solutions. In accordance with variouscommunications standards, a single-sideband RF output is often requiredat the antenna (e.g., antenna array).

In a first transmitter architecture, signals in quadrature (e.g., havingan in-phase signal and a quadrature signal in a quadrature phaserelationship to the in-phase signal) at baseband frequencies are mixedwith stages of quadrature local oscillator signals (e.g., for generatingsingle sideband RF transmit signals for transmission). The firsttransmitter architecture is typically suitable for integratedsystem-on-chip implementation.

In a second transmitter architecture, the transmitter generates thesignals in quadrature by phase shifting band-limited local oscillatoranalog signals (e.g., where the single sideband RF transmit signals fortransmission are generated by one or more mixer stages, where each mixerstage is responsive to local oscillator signals in quadrature, where thelocal oscillator signals in quadrature are generated in accordance witha selected frequency).

The first and second transmitter architectures are applicable torespective first and second receiver architectures where, for example,the output of the down-conversion RF circuitry can be a differentialsignal (in accordance with the first receiver embodiment) or quadraturedifferential signals at analog/baseband frequency levels (in accordancewith the second receiver embodiment).

Various transmitter architectures using mixers for generating outputsignals are responsive to quadrature input signals and quadrature localoscillator (LO) signals in accordance with:cos(ω_(LO)+ω_(BB))t=cos(ω_(LO) t)cos(ω_(BB) t)−sin(ω_(LO) t)sin(ω_(BB)t)  (Eq. 1)where ω_(BB) is a first quadrature input signal at a baseband frequency,ω_(LO) is a second quadrature input signal having a frequency higherthan a baseband frequency and t is time. The first and second quadratureinput signals each include a first signal and a second signal having thefrequency of the included first signal and having a 90-degree phaserelationship with the included first signal.

FIG. 1A is a schematic diagram of an architecture of a beam-focusingMIMO transmitter 100. The digital signal processing element 101 isarranged for back-end signal processing (e.g., processing of signals fortransmitting and processing of digitized received signals). The digitalsignal processing element 101 generates transmit data and phaseinformation for transmitting a selected output signal. The transmit dataand phase information is transferred to a number M of quadrature DACs(to digital-to-analog converters 102-A1 and 102-A2 for a first channeland to a sequence of (e.g., the selected number of) M channels extendingto DACs 102-M1 and 102-M2 for the last channel). The number M ofbaseband filters (103-A1 and 103-A2 and a sequence of baseband filtersextending to 103-M1 and 103-M2) are arranged for reducing aliasing andharmonics of the respective baseband signal generated by the respectivequadrature digital-to-analog converters (e.g., DACs 102-A1 and 102-A2through 102-M1 and 102-M2). After the baseband signal is filtered by theanti-aliasing filters (e.g., filters 103-A1 and 103-A2 through 103-M1and 103-M2), the filtered baseband signals are respectively up-convertedusing a number M of up-converting quadrature mixers (mixers 104-A1 and104-A2 through 104-M1 and 104-M2), which output a respectiveup-converted signal coupled to an input of a number M of the highfrequency power amplifiers (106-A and the sequence of amplifiersextending to 106-M). The outputs of the high frequency power amplifiersare respectively coupled to each the antennas (107-A and the sequence ofantennas extending to 107-M). The beam-focusing MIMO transmitter 100includes a phase shifter 108 for generating LO signals and quadrature.

FIG. 1B is a schematic diagram of various example embodiments of a phaseshifter 108 for generating LO signals and quadrature. For example, phaseshifter 108 can be a phase shifter in accordance with any of 108-A,108-B, and 108-C phase shifter embodiments.

The phase shifter 108-A includes a number M voltage-controlledoscillators (VCOs 109-A and the sequence of VCOs extending to 109-M),where each of the M number of VCOs (e.g., 109) includes an outputcoupled to a respective set of inputs of each of the quadrature (QUAD)dividers (110-A and the sequence of dividers extending to 110-M). Thephase shifter 108-A embodiment is suitable (for example) for VCOs whereeach such VCO covers a narrow tuning range, and where many such VCOs(e.g., where each of the VCOs covers a different frequency range of alarger frequency range) are used to cover an entire frequency range ofcoverage (e.g., by selecting a particular VCO capable of operating at atarget frequency within the frequency range of coverage). The respectiveoutputs (e.g., LO-I and LO-Q) of each of the quadrature dividers (e.g.,110-A through 110-M) are output to a respective mixer in accordance with(“<”) signals φ₁ through φM. The quadrature divider outputs aredifferentially coupled to respective up-converting mixers (e.g., arespective pair of the up-converting mixers 104).

The phase shifter 108-B shows a configuration where the oscillators(111-A through 111-M) are multiplexed via a multiplexer 112, which inturn selectively outputs a selected baseband signal generated by aselected one of a number M VCOs (111-A and the sequence of VCOs to111-M) and provided the selected output to the number M of quadraturedividers (113-A and the sequence of dividers to 113-M) to obtain thequadrature outputs for respectively coupling to a respective pair of thenumber of M mixers (e.g., up-converting mixers) 104.

The phase shifter 108-C shows a configuration where the output of a VCO114 is provided to a number M of quadrature dividers (115-A through115-M), where each quadrature divider 115 provides quadrature outputscoupled to a respective M quadrature up-conversion mixer. Accordingly,the phase shifter 108-C is suitable for receiving an input signal from awide tuning range VCO.

FIG. 2 is a schematic diagram of an architecture of a beam-focusing MIMOreceiver. A multipath receiver 200 includes an array of cooperatingreceiver elements. A number N of front-end antennas (206-A through206-N) are respectively coupled to a number N of low noise amplifiers(LNAs 205-A through 205-N). Outputs from the LNAs are respectivelycoupled to a respective differential input of a corresponding number Nof low-pass baseband filters (filters 203-A1 and 203-A2 through 203-N1and 203-N2) respectively. The outputs from the low-pass baseband filters(e.g., 203) are respectively coupled in quadrature to respective pairsof the number N of the analog-to-digital converters (ADCs 202-A1 and202-A2 through 202-N1 and 202-N2). The outputs of the analog-to-digitalconverters (e.g., ADCs 202) are coupled to the receiver digital signalprocessor 201, where the DSP 201 is arranged to process each respectivedigital output stream such that, for example, the directional lobes ofthe front-end antennas are effectively oriented by software control(which, for example, enhances directional gain and increasessignal-to-noise ratios).

FIG. 3 is a schematic diagram of VCO polyphase quadrature signalgeneration circuits. For example, VCO polyphase quadrature signalgeneration circuits can be VCO polyphase quadrature signal generationcircuits in accordance with any of 300-A, 300-B, and 300-C.

Quadrature generator 300-A is a single-VCO polyphase quadrature signalgenerator and includes a differential VCO 301 for coupling a (e.g.,in-phase) differential signal to a buffer 302, which in turn drives(e.g., using in-phase differential signaling) the polyphase quadraturegenerator (PPF I/Q GEN) 303. The polyphase quadrature generator 303 isarranged to generate output pairs for each of the in-phase signal I andquadrature signal Q, which are respectively coupled to a number M ofphase shifters 304-A through 304-M, where each phase shifter (e.g., 304)is arranged to phase shift a respective coupled input signal pair. Eachphase shifter 304 is arranged to operate in accordance with the phaseshift operations:cos(ω_(LO) t−φ _(x))=α cos(ω_(LO) t)+β sin(ω_(LO) t), wheretan(φ_(x))=β/α  (Eq. 2)sin(ω_(LO) t−φ _(x))=α sin(ω_(LO) t)−β cos(ω_(LO) t)  (Eq. 3)where φ_(x) is a selected phase shift for each transmitter (or receiver)element. The parameters α and β are individually controllable viasoftware (e.g., under control of software executing on DSP 201) for eachphase shifter (e.g., 304-A) and are selected to control the phase shiftprovided by an individual phase shifter (e.g., 304-A). Each phaseshifter in a transmitter element is programmed with a potentiallydifferent value for α and β, where α_(i)≠α_(j), and β_(i)≠β_(j), forexample, where i and j indicate a particular adjacent transmitterelements.

The differential output of each phase shifter (e.g. one of 304) isrespectively coupled to a respective differential input of one of anumber M of individual quadrature generators (305-A through 305-M). Theoutput pair (which includes in-phase “I” and quadrature-phase “Q”signals) of each quadrature generator (e.g., one of 305) is coupled tothe quadrature and in-phase inputs of a respective up-converting mixer(e.g., 104 or 204). The quadrature generator 300-A consumes relativelylarge amounts of power and area, in part based on the number of buffers302 for compensating the signal loss associated with the polyphase phaseshifter networks. Further, the buffers 302 also increase out-of-bandphase noise based on the associated thermal noise contribution.

Quadrature generator 300-B is a dual-VCO quadrature signal generator inaccordance with the quadrature signal generator method introduced above.In general, quadrature generator 300-B includes two VCOs each of whichincludes cross-coupled outputs for generating quadrature signals. Thequadrature generator 300-B includes two cross-coupled VCOs 306-A (VCOI)and 306-B (VCOQ). The VCOs 306-A and 306-B provide quadrature outputs,which can be vector summed using a number M of vector modulators (e.g.,vector modulators 307-A through 307-M) for phase shifting input signalsin accordance with Eq. 2 and Eq. 4. The outputs of the vector modulators(e.g., 307) are coupled to respective inputs of M cooperating (e.g.,concurrently working together) quadrature phase shifters 308-A through308-M. Each of the quadrature phase shifters (e.g., 308) includesdifferential quadrature outputs, which are coupled to respective inputsof a respective transmitter mixer (e.g., one of the up-converting mixers104). The quadrature generator 300-B occupies a relatively large area(e.g., often consumed by the VCO inductors) and entails increasedcross-talk resulting from included inductive components.

Quadrature generator 300-C is a single-VCO passive-quadraturephase-shifting generator in accordance with the single-VCOpassive-quadrature phase-shifting method introduced above. In general,quadrature generator 300-C includes a single VCO 308, which includes a(e.g., differential) output coupled to a quadrature hybrid (Q-HYB) 309(e.g., in accordance with transmission line theory). The differentialoutputs of the quadrature hybrid 309 can be vector summed using a numberM of vector modulators (310-A through 310-M) for phase shifting inputsignals in accordance with Eq. 2. The outputs of each of the vectormodulators (e.g., 310) are coupled to a respective input of one of thenumber of M concurrently working quadrature phase shifters (311-Athrough 311-M). Each of the quadrature phase shifters (e.g., 311)includes differential quadrature outputs, which are coupled torespective inputs of a respective transmitter mixer (e.g., one of theup-converting mixers 104). The quadrature phase shifters (e.g., 311) canbe implemented using quadrature hybrid manufacturing technology.

In a transceiver architecture, phase shifting can be provided by: (a)baseband or low frequency transceiver circuitry, (b) LO or signalgeneration frequency transceiver circuitry, and/or (c) RF (e.g.,transmit/receive) circuitry. In a transmitter architecture, phaseshifting can be performed in baseband or LO frequency circuitry because(for example) a single sideband combination is determined in response tothe baseband or LO frequencies. Performing phase shifting at RFfrequency consumes power and occupies area of a substrate. In a receiverarchitecture, phase shifting can be performed in RF or LO frequencycircuitry. Generally, phase shifting is performed using circuitryoperating at LO frequencies in part due to the higher phase accuraciesobtained in the signal generation, as well as less noise contributioncompared to the phase shifter operation in the signal path (operating inlow frequency baseband or RF bands).

FIG. 4 is a schematic diagram of an architecture of a transmitterelements of a beam-focusing MIMO transmitter including cross-coupledquadrature mixer elements in accordance with example embodiments. TheMIMO transmitter includes a number N of transmitter elements (400-Athrough 400-N), where each transmitter element works in parallel (andconcurrently) with other such transmitter elements to generate outputsignals for combining spatially when radiated by the respectiveantennas. Accordingly, each of the N transmitter elements 400-A through400-N cooperate as elements of a MIMO transmitter arranged fortransmitting output signals for combining spatially (e.g., for improvingeffective power and directivity by constructive interference ofspatially combined transmitted waveforms). The baseband functionality ofeach transmitter element (e.g., 400-A) can be implemented usingintegrated circuit technologies different from the integrated circuittechnologies used to implement the RF functionality (e.g., basebandcircuit portions can be manufactured using CMOS technologies and RFcircuit portions can be manufactured using silicon germanium).

Each of the transmitter elements includes a number N of polyphase phaseshifters (e.g., PPFs 401-A through 401-N) at the input of eachtransmitter element (e.g., one of transmitter elements 400). Thepolyphase phase shifters (e.g., 401-A to 401-N) can use differentialinputs (e.g., coupled from the outputs of a buffer 302) or quadratureinputs input (e.g., coupled from the outputs of a polyphase quadraturegenerator 303) in accordance with a particular system application. Theinputs of the polyphase phase shifter are arranged to receive a basebandinput information signal (BB′) generated by, for example, a DACprogrammed by processor in accordance with information to betransmitted. When the baseband input is differential, the polyphasephase shifter (e.g., one of PPFs 401) processes the baseband signal byquadrature phase shifting and filtering. The degree of phase shift canoptionally be formed as “hard-wired” and/or “selectable alternatives(e.g., programmable switches activated) after deployment” using analogand/or digital techniques.

The polyphase phase shifters (e.g., PPF 401-A) are arranged as multiplestages of low pass and high pass filters, having low pass and high passcutoffs determined as single-order time-constants. In some embodiments,the polyphase phase shifters are arranged 1) to obtain quadrature phasesof a given waveform, and 2) to perform phase shifting. The degree ofphase shifting is determined by a vector summation of orthogonal phasesusing a scalar factor (parameters α and β). For example, the phase shiftis determined in accordance with φ(t)=α*I(t)+β*Q(t), which is dependentof the I(t) and Q(t) provided by the quadrature signals, which arerespectively attenuated by the parameters α and β.

Outputs of the polyphase phase shifters 401 are coupled to thetrigonometric weighting scalers (e.g., 402-A1 and 402-A2 through 402-N1and 402-N2). Each of the trigonometric scalers (e.g., one of 402)provides an attenuated version of the input signal in accordance withthe trigonometric scaling factors α and β. The attenuation factors aredetermined for one or more design applications (e.g., at boot time bythe DSP 101 or initially by a circuit designer) based on the degree ofphase shift to be performed by the transmitter element (e.g., 400-A) inaccordance with Eq. 2 and Eq. 3. The outputs of the trigonometricweighting scalers 402 are coupled to a first mixer stage 410 ofquadrature mixer elements (e.g., 403-A1, 403-A2, 403-A3, 403-A4 through403-N1, 403-N2, 403-N3, 403-N4).

The output currents from the quadrature mixer elements 403-A1 and 403-A3are summed together using (e.g., wire-implemented) combiner 404-A1 toobtain a first phase shifted version of input baseband signalup-converted to an IF frequency in accordance to Eq. 2. Similarly, theoutput currents from the quadrature mixer elements 403-A2 and 403-A4 aresubtracted from each other using (e.g., wire-implemented) combiner404-A2 to obtain a second phase shifted version of input baseband signalup-converted to an IF frequency in accordance with Eq. 3.

Accordingly, the outputs of combiners 404-A1 and 404-A2 are quadraturephase signals (signals in quadrature) at an IF frequency where the IFfrequency is the sum of the baseband frequency and the first localoscillator frequency (e.g., IF=LO1+BB). The output of combiner 404-A1 isgenerated in accordance with αI+βQ (see Eq. 2) whereas the output ofcombiner 404-A2 is generated in accordance with αQ−βI (see Eq. 3). Theoutput of the combiners 404-A1 and 404-A2 (e.g., which is the quadratureIF signal) are coupled to a second mixer stage 420 of quadrature mixers405-A1 and 405-A2. Generally, the outputs of the first mixer stage 410of quadrature mixers are respectively coupled to respective inputs ofthe second mixer stage 420 of quadrature mixers (which operate inaccordance with LO signals different from the LO signals of the firstmixer stage 410).

Expanding Eq. 1 in accordance with at least two quadrature stages of theLO signal:cos(ω_(LO1)+ω_(LO2)+ω_(BB))t=cos(ω_(LO1) t)cos(ω_(LO2) t)cos(ω_(BB)t)−sin(ω_(LO1) t)sin(ω_(LO2) t)cos(ω_(BB) t)−sin(ω_(LO1) t)cos(ω_(LO2)t)sin(ω_(BB) t)−cos(ω_(LO1) t)sin(ω_(LO2) t)sin(ω_(BB) t)  (Eq. 4)where ω_(LO1) is a second quadrature input signal having a frequencyhigher than the baseband frequency and ω_(LO2) is a third quadratureinput signal having a frequency higher than the baseband frequency anddifferent from the frequency of ω_(LO1). In various embodiments, theexample basic single-sideband combination of quadrature phases at eachof the constituent signals shown in (Eq. 1) and (Eq. 4) above can beextended to include multiple stages of quadrature input signals (e.g.,including more than three frequencies).

A single-sideband output signal is generated in response to quadraturephasing for each of the quadrature input signals. For example: for asingle-stage up-conversion, the baseband and LO frequencies are eachprovided in quadrature; and for a two-stage up-conversion, each of thethree components (baseband, first LO and second LO) are provided inquadrature.

The quadrature generation at baseband frequencies are often implementedusing analog circuit techniques cost-effective for low frequencies,while the quadrature generation circuitry at higher LO frequencies aregenerally more expensive (e.g., including in terms of both power andarea). Additionally, the circuitry for quadrature generation at bothbaseband frequency levels and LO frequency levels are relativelyspecific for selected frequencies. For a single design to work acrossvarious selected frequencies, additional circuitry is included as areusable transceiver circuit design to accommodate changes in frequencyplans (e.g., in accordance with different frequency usage assignment indifferent countries), for which the transceiver circuit is to be used incell phones of different carriers. For example, a transmitter designedto operate at both 28 GHz (e.g., in the United States and Europe) and 35GHz (e.g., in Japan) can consume relatively large substrate areas due toquadrature hybrids arranged to operate at a selected one of the twofrequencies.

Each of the transmitter elements 400-A through 400-N are programmed inaccordance with different values of α and β, leading to the requiredphase granularity (e.g., of 180/N) of the beam-focusing transmitterarray. Accordingly, the trigonometric weighting scalers 402 are forphase shifting the output information signal (e.g., generated inresponse to an input information signal received from a processor forencoding and transmission) of the transmitter element in accordance witha phase shift selected by the system DSP for spatial combination (e.g.,through constructive combination) of the transmitted waveforms ofadjacent transmitter elements.

As discussed below with respect to FIG. 9, trigonometric scaling isoptionally implemented using a factor for scaling the mixers. The mixerscan be scaled, for example, by segmenting a baseband transistor fordriving an associated up-conversion combiner 404 into functional smallersegments and selecting one or more of the functional smaller segmentedtransistors. The baseband transistor can be segmented by parallelingsmaller transistors and selecting (e.g., via mask programming orprogrammable transistor switches) various ones of smaller transistorssuch that the selected transistors scale in accordance with a selectedtrigonometric ratio.

Before amplification and transmission, the first and second LO stagefrequencies (e.g., LO1 and LO2) are both generated in quadrature and arereceived from the LO generation blocks 500-A and 500-B (described belowwith respect to FIG. 5).

Combiner elements of the up-conversion combiners (e.g., 404-A1 and404-A2 through 404-N1 and 404-N2) are arranged to combine input signalsin (e.g., an electrical) current mode by, for example, connecting twowires together without additional components. The wired combinerconnections save both power, area, and avoids linearity limitationsbecause of no additional active circuit components existing in theassociated signal path.

The baseband shifter (e.g., 401), the scalar amplifier (e.g., 402), thefilter (e.g., 403), and an up-combiner (e.g., 404) of a transmitterelement (e.g., 400-A) are arranged for vector modulations. Accordingly,baseband phase shifters (e.g., one of 401), scalar amplifiers (e.g., anassociated one of 402), filters (e.g., an associated one of 403), and anup-converter (e.g., current summing) mixers (e.g., an associated one of404) are vector modulators. The outputs of the vector modulators arerespectively coupled to the second stage of the respective mixers (e.g.,405-A and 405-B to 405-N and 405-N). The second stage of mixers 405receives a second set of quadrature signals (e.g., LO_(2I), LO_(2Q))from the LO signal generation elements (e.g., 500-A or 500-B).

Outputs from the second stage of the mixers (e.g., 405-A and 405-B) ineach transmitter element are coupled to a respective combiner (e.g.,current summer 406-A through 406-N) to generate a single sidebandoutput. The current summers are combiners implemented as a (e.g., wired)connection (which does not require any active circuitry, conservescircuit power and area, and avoids signal distortion, which helpspreserve the dynamic range of the signal being summed). Output signalsfrom the current summers (e.g., 406-A through 406-N) are coupled torespective power amplifiers (e.g., 407-A through 407-N), which areselected for providing amplification of respective output signals at aselected center frequency of interest.

In an embodiment, the current summers 406 and the power amplifiers 407can be eliminated, and the current summation (otherwise performed by thecurrent summers 406) can occur at the antenna by coupling the outputs ofa pair of up-converting mixers (e.g., combiners 404-A and 404-B havingcross-coupled inputs) directly together at the antenna. Coupling theoutputs of the respective pairs of up-converting combiners at theantenna provides relatively high degrees of linearity when, for example,coupling the combiners (such as discussed below with reference to FIG. 9and FIG. 10) to inductively loading antennas, which can be configuredfor near-field communication.

Cost-effective, next-generation MIMO systems optionally include thecircuitry to reconfigure transmitter elements to different operatingfrequencies such that, for example, the same mobile hardware circuitscan be used in multiple countries over the world (e.g., whereinindividual countries might allocate frequency “spectrum” differentlyfrom other countries). Two commonly used frequency bands include RFcenter frequencies of 28 GHz and 35 GHz. Accordingly, reconfigurabletransmitter and receiver elements are described herein for operating inthese bands. (Some of the front-end tuning elements have off-substratecomponents, which can be relatively easily changed, and whichfacilitates adaptation for different antenna dimensions and structuresused for optimum performance.)

Two embodiments for quadrature signal generation at a given frequency ofinterest are discussed. A first embodiment includes a quadraturegeneration for a sinusoidal tone, which can be the output of the VCO.The output of the VCO can be coupled as the LO signal, where the LO is arelatively low frequency signal such that VCO can be relatively easilytuned in accordance with a specified transmitter frequency. Astransmitter frequencies changes from 28 GHz to 35 GHz, a wide tuningrange VCO and a wideband quadrature phase shifter can be used toaccommodate the change in frequency. In a second embodiment, a differentset of tuning elements (e.g., resonators for VCO and phase shifterelements for the quadrature) can be formed by changing metal masksduring integrated circuit manufacture of a circuit in accordance withthe second embodiment.

In both embodiments a processor (such as a DSP) is arranged to programand/or reprogram the local oscillator (e.g., multiplication orderoperation units 500-A or 500-B discussed below, which are arranged toperform multiplication order operations such as multiplication anddivision in response to selected integers) for generating first andsecond LO signals in accordance with a first carrier frequency and forgenerating first and second LO signals in accordance with a secondcarrier frequency. (The DSP is also arranged to program variouspolyphase phase shifters, trigonometric weighting scalers, selectingtransistor segments of multi-segment transistors of mixers—see FIG. 9and FIG. 10, for example—and other frequency-dependent configurablecircuitry.)

Various methods for generating a LO signal at high frequencies include:(a) a frequency-doubled method in which a VCO operating at twice the(e.g. transmitter) frequency is further divided by a factor of 2 (e.g.,such that the doubled frequency is divided by 4); (b) a single-VCOpolyphase-quadrature phase-shifting method in which a VCO operates atthe same frequency as the transmitter, but is phase shifted forbeam-focusing operations in response to polyphase quadrature generation;(c) a dual-VCO method in which two VCOs are cross-coupled with eachother to generate quadrature signals and are phase shifted forbeam-focusing operations in response to polyphase quadrature generation;and (d) a single-VCO passive-quadrature phase-shifting method in which asingle VCO is phase shifted for beam-focusing operations in response tobeing coupled to a passive quadrature hybrid for generating quadraturesignals, where the generated quadrature signals are phase shifted forthe beam-focusing transmitter array.

The frequency-doubled method described above for generating a LO signalhigh frequencies, for example, includes a VCO having a relatively widetuning range (at least plus or minus 20 percent of a center frequencyoutput), where the VCO is arranged to operate at 56 GHz and 70 GHzrespectively for generating output frequencies of either 28 GHz or 35GHz. The frequency-doubled method entails relatively large amounts oflayout space and power (for example, multiple VCOs can be provided,where each VCO has a different frequency range such that a widefrequency tuning range is accommodated by selecting a different VCO inaccordance with a selected transmission frequency).

FIG. 5 is a schematic diagram of local oscillators (LO) for modulatingtransmitter elements of a beam-focusing MIMO transmitter in accordancewith example embodiments. In general, a local oscillator includes asingle phase locked loop (PLL) VCO for driving for frequency dividersand multipliers. The signal generation path is arranged as a fullydifferential architecture, which lessens transmission line effectsassociated with power and ground lines. A fully differential design isgenerally more immune to linearly translated common-mode cross-talkinduced by, for example, fluctuations in voltages in structures and/orbusses of the power supply, ground, and/or substrate. Accordingly, thesignal distribution in the described embodiment is fully differential(e.g., starting from the output of a differential VCO and continuing tothe output of a mixer interface).

In various embodiments, the local oscillator includes a VCO forgenerating differential signals as a master frequency signal. The masterfrequency signal is coupled differentially to a multiplication-orderoperation unit, which is arranged to generate a first derived frequencysignal (e.g., a first LO output signal) in response to the masterfrequency signal in accordance with a first multiplication-orderoperation (e.g., either division or multiplication), and for generatinga second derived frequency signal (e.g., a second LO output signal) inresponse to the first derived frequency signal in accordance with asecond multiplication-order operation (e.g., either multiplication ordivision), wherein the second multiplication-order operation is amultiplication-order operation that is inverse to the firstmultiplication-order operation. In the single-oscillator embodiment andthe dual-oscillator embodiment discussed below, the multiplication-orderoperation unit executes integer frequency division and integer frequencymultiplication operations pon of respective signal inputs.

In a single-oscillator embodiment, the LO 500-A includes a VCO 501 forgenerating the master frequency signal. The VCO 501 master frequencysignal is coupled to an even-integer divider 502. The even-order divider502 divides the VCO frequency by an even integer P. For simplicity ofthe embodiment, P can be a number derived from a cascade ofdivide-by-two circuits (e.g., modulo-2 division). The even-order divider502 can include a cascade of even/odd dividers coupled with adivide-by-2 divider (e.g., such that the input frequency is divided byan even number). The even-order divider 502 outputs an evenly dividedfrequency as a fifty-percent duty signal in differential quadraturephases.

The even-order divider 502 output is coupled to an odd-order frequencymultiplier 503, which multiplies the input frequency by an integer Q.The frequency multiplier 503 is typically implemented using aninjection-locked frequency multiplier or an overdriven multiplier forconverting energy from fundamental frequency to odd-order harmonics.Accordingly, the two frequencies generated from this LO generationsystem are given by:LO_(1I)=LO/P<0°, LO_(1Q)=LO/P<90°  (Eq. 5)LO_(2I)=LOQ/P<0°, LO_(2Q)=LOQ/P<90°  (Eq. 6)where P is the even-divisor integer, Q is the odd-multiplier integer,and the “<” operator indicates the phase of the associated signal.

In accordance with Eq. 5 and Eq. 6, some embodiments of theup-conversion mixer elements in the transmitter array are describedbelow as (A), (B), and (C).

(A) Up-conversion mixers are arranged to operate responsive to signalsfrom a single LO (e.g., LO2) in accordance with:RF=LOQ/P=>LO=P/QRF  (Eq. 7)such that the VCO can be operated at a frequency lower than the RFfrequency. When P=2 and Q=3, the VCO can operate at two-thirds of the RFfrequency. For a 35 GHz RF carrier frequency, a VCO can operate at acenter frequency of 24 GHz. Similarly, when P=2 and when Q=5, the VCOcan operate at a frequency of two-fifths of the RF frequency. For a 35GHz RF carrier frequency, a VCO can operate a center frequency of 14GHz. The selection of the frequency multiplier and dividers achievesgreater frequency tuning ranges, while consuming lower power using even(e.g., digital divide-by-2) dividers.

(B) Up-conversion mixers are arranged to operate responsive to signalscombined from a pair of local oscillators (e.g., LO1 and LO2) in anadditive manner. For example, the upper sideband is generated in part bya first mixer (e.g., 403-A1 and 403-A2 collectively) and the uppersideband is generated in part by a second mixer (e.g., 405-A1 and 405-A2collectively). Alternatively, the lower sideband can be generated inpart by the first mixer (e.g., 403-A1 and 403-A2 collectively) and inpart by the upper sideband of the second mixer (e.g., 405-A1 and 405-A2collectively).

In like manner, a two step up-conversion can also be performed in anadditive manner (e.g., where the upper sideband is coupled as inputs forboth stages of mixing). The up-conversion mixers are arranged to operateresponsive to signals combined from a pair of local oscillators in anadditive manner in accordance with:

$\begin{matrix}{{RF} = {{{LO}_{1} + {LO}_{2}} = {{{LO}\frac{Q + 1}{P}\mspace{14mu}\text{=>}\mspace{14mu}{LO}} = {\frac{P}{Q + 1}{RF}}}}} & \left( {{Eq}.\mspace{14mu} 8} \right)\end{matrix}$

For a given RF center frequency of 35 GHz and the values P=2 and Q=3 inan example, the VCO center operating frequency is 17.5 GHz, which is asub-harmonic of the RF carrier frequency. Accordingly the architectureis (e.g., mathematically) immune to carrier frequency drift resultingfrom effects from parasitic coupling in responsive to high poweramplifier (PA) array output power (e.g., because the parasitic inputsare direct multiples of the VCO output frequency).

Up-conversion mixers are arranged to operate responsive to signalscombined from a pair of local oscillators (e.g., LO1 and LO2) in asubtractive manner (whereas swapping connections on the mixer inputswould cause the signals to be combined in an additive manner). Forexample, the upper sideband is generated by a first mixer (e.g., 403-A1and 403-A2 collectively) and the upper sideband is generated by a secondmixer (e.g., 405-A1 and 405-A2 collectively). Alternatively, the lowersideband can be generated by the first mixer (e.g., 403-A1 and 403-A2,collectively) and the upper sideband of the second mixer (e.g., 405-A1and 405-A2 collectively).

In the embodiment, the single sideband transmitter TX can be realizedusing a single step up-conversion circuit, which includes for exampleLO_(2I) and LO_(2Q). For a direct up-conversion architecture, thefrequencies from LO_(2I) and LO2 _(Q) are combined in a subtractivemanner (e.g., upper sideband generation is performed by the first mixer403-A1, -A2 and lower sideband generation is performed by the secondmixer 405-A1, -A2):

$\begin{matrix}{{RF} = {{{LO}_{2} - {LO}_{1}} = {{{LO}\frac{Q - 1}{P}\mspace{14mu}\text{=>}\mspace{14mu}{LO}} = {\frac{P}{Q - 1}{RF}}}}} & \left( {{Eq}.\mspace{14mu} 9} \right)\end{matrix}$

Accordingly, when P=2, Q=5, and the RF center frequency is 35 GHz, theVCO frequency becomes 17.5 GHz, which is a sub-harmonic of the RFcarrier frequency, such that the architecture is (e.g., mathematically)immune to frequency deviations from high PA output power via variouscoupling mechanisms.

Vector diagrams 500-C and 500-D represent vectors of the output offrequency multipliers when a frequency multiplier receives a signal inquadrature. Vector diagram 500-C shows input vectors of 0 degrees and+90 degrees phases for I and Q (where the y-component Q ispositive-negative-positive for the fundamental, third, and fifthharmonics, respectively). In contrast, vector diagram 500-D shows thevectors generated when the input signals are 0 and −90 degree phases forI and Q (where the y-component Q is negative-positive-negative for thefundamental, third, and fifth harmonics, respectively).

Because of the phasor rotation of the quadrature signal, the polarity ofthe quadrature signal at harmonics of input signal alternates (e.g.,between +90 and −90 degrees) while the in-phase signal maintains thesame vector position (e.g., 0 degrees). For an in-phase signal (e.g.,where the I signal maintains a zero degree phase), the odd-orderharmonics are of the same phasor (e.g., where the third and fifthharmonic signals are also of zero degree phase).

In contrast, the odd-order harmonics of a quadrature signal alternatepolarities (e.g., change by 180 degrees) between each successive oddharmonic. For example, the vector diagram 500-C shows a quadrature phaseof +90 degrees for the first (1X) harmonic, a quadrature phase of −90(e.g., +270) degrees for the third harmonic, and a quadrature phase of+90 degrees for the fifth harmonic (e.g., which is the same quadraturephase angle of the first harmonic). Also for example, the vector diagram500-D shows a quadrature phase of −90 degrees for the first (1X)harmonic, a quadrature phase of +90 degrees for the third (3X) harmonic,and a quadrature phase of −90 degrees for the fifth (5X) harmonic (e.g.,which is the same quadrature phase angle of the first harmonic).

In an embodiment, the quadrature phase signals are inverted formultiplication of the third order harmonic after which the multipliedinverted harmonics are combined with (e.g., non-inverted) multipliedharmonics to generate a homogenous quadrature signal where the odd-orderharmonics (e.g., at least the first and third) are of the samequadrature phase signal. The homogenous quadrature signal is coupled tothe up-conversion mixers as the Q signal.

In an alternate embodiment, the quadrature phase signals are invertedfor multiplication of the first and fifth order harmonics and combinedwith an (e.g., non-inverted) third order harmonic to generate thehomogenous quadrature signal. In a fully differential embodiment, thedissimilar harmonics are inverted by reverse coupling of wires, suchthat (for example) additional power and layout area are not necessarilyconsumed.

In a dual oscillator embodiment, the LO 500-B includes dual VCOs 504,with the output of each VCO 504 coupled to a respective input of thequadrature amplifier 505. The dual VCOs 504 are cross-coupled to ensurecommon phase integrity. Accordingly, LO 500-B provides is similar to LO500-A in that, for example, multiply operations are performed first anddivisions are performed later. In the dual-oscillator embodiment, aquadrature VCO 504 is used to obtain the differential quadrature signalsat a selected LO frequency. The quadrature VCOs 504 are coupled toquadrature multiplier 505, which is arranged to multiply the inputfrequency by amplifying (and homogenizing) odd-harmonic energy tomultiply the input frequencies in accordance with the odd harmonics ofthe input frequencies. The homogenous multiplied frequency (inquadrature) is output by the quadrature multiplier 505 and is furthercoupled with even divider P 506. The outputs of the quadraturemultiplier 505 are coupled and applied in a manner similar to the outputof the frequency multiplier 503 described above. The LO 500-A and the LO500-b can be applied to at least four embodiments described below as(A), (B), (C), and (D).

(A) A first embodiment includes single sideband up-conversion mixersusing LO1 signals (e.g., only), where the quantities P and Q aremutually exchanged (e.g., swapped):RF=LOQ=>LO=1/QRF  (Eq. 10)

Accordingly, when Q=3 and the RF center frequency is 35 GHz, the VCOfrequency becomes 12.67 GHz, which is a sub-harmonic of the RF carrierfrequency, such that the architecture is (e.g., mathematically) immuneto frequency deviations from high PA output power via various couplingmechanisms.

(B) A second embodiment includes single-sideband up-conversion mixersusing LO2 signals (e.g., only), where the quantities P and Q aremutually exchanged (e.g., swapped):RF=LOP/Q=>LO=Q/PRF  (Eq. 11)

Accordingly, when P=6, Q=3 and the RF center frequency is 35 GHz, theVCO frequency becomes 17.5 GHz, which is a sub-harmonic of the RFcarrier frequency, such that the architecture is (e.g., mathematically)immune to frequency deviations from high PA output power via variouscoupling mechanisms.

(C) A third embodiment includes single-sideband up-conversion mixersusing a combination of LO1 and LO2 in additive manner:

$\begin{matrix}{{RF} = {{{LO}_{1} + {LO}_{2}} = {{{LO}\frac{Q\left( {P + 1} \right)}{P}\mspace{14mu}\text{=>}\mspace{14mu}{LO}} = {\frac{P}{Q\left( {P + 1} \right)}{RF}}}}} & \left( {{Eq}.\mspace{14mu} 12} \right)\end{matrix}$

Accordingly, when P=2, Q=3 and the RF center frequency is 35 GHz, theVCO frequency becomes 7.8 GHz, which is a sub-harmonic of the RF carrierfrequency, such that the architecture is (e.g., mathematically) immuneto frequency deviations from high PA output power via various couplingmechanisms. Note that Q=1 leads to a simple case of sliding IFarchitecture (where the ratio of the LO₁/LO₂ determines a division ratioin quadrature over the frequencies used in single sidebandcommunications)

(D) A fourth embodiment includes single sideband up-conversion mixersusing a combination of LO1 and LO2 in subtractive manner:

$\begin{matrix}{{RF} = {{{LO}_{1} + {LO}_{2}} = {{{LO}\frac{Q\left( {P + 1} \right)}{P}\mspace{14mu}\text{=>}\mspace{14mu}{LO}} = {\frac{P}{Q\left( {P - 1} \right)}{RF}}}}} & \left( {{Eq}.\mspace{14mu} 13} \right)\end{matrix}$

Accordingly, when P=2, Q=3 and the RF center frequency is 35 GHz, theVCO frequency becomes 12.67 GHz, which is a sub-harmonic of the RFcarrier frequency, such that the architecture is (e.g., mathematically)immune to frequency deviations from high PA output power via variouscoupling mechanisms. When P=4 and Q=3, the VCO center frequency becomes15.6 GHz. From the two embodiments of quadrature LO signal generationschemes (e.g., LO 500-A and LO 500-B) described above, more flexibilitycan be obtained by having a programmable even order divider and aprogrammable odd-order multiplier (e.g., where P and/or Q areprogrammable by software executing on a processor). Similarly, the sameVCO can be used to generate output frequencies over multiple selectedfrequency bands by generating a combination of a first sideband (upperor lower) for a first RF frequency band (e.g. 28 GHz), and a secondsideband (lower or upper) for a second RF frequency band (e.g. 35 GHz).In order to achieve a smaller tuning range for the VCO, the secondsideband is a sideband that is complementary to the first sideband(e.g., the first sideband is an upper sideband and the second sidebandis the lower sideband). In accordance with the description herein, thetuning range of the VCO is reduced by using complementary sidebandinjection with respect to the VCO programmed to operate at a firstoperating frequency and a second operating frequency.

In a single mixing stage example using LO 500-A, where P=2, Q=3, andIF=2.5G and for a first carrier frequency of 28 GHz, a VCO generating amaster frequency 17 GHz is indicated when the in-phase andquadrature-phase mixers are programmed for low-side injection (e.g., ofthe upper sideband). When the in-phase and quadrature-phase mixers areprogrammed for high-side injection (e.g., of the lower sideband), a VCOgenerating a master frequency 20.33 GHz (result A) is indicated foroperating (assuming P=2, Q=3, and IF=2.5G and the first carrierfrequency RF=28 GHz).

In the single mixing stage example using LO 500-A, where P=2, Q=3, andIF=2.5G and for a second carrier frequency of 35 GHz, a VCO generating amaster frequency 21.67 GHz (result B) is indicated when the in-phase andquadrature-phase mixers are programmed for low-side injection (e.g., ofthe upper sideband). When the in-phase and quadrature-phase mixers areprogrammed for high-side injection (e.g., of the lower sideband), a VCOgenerating a master frequency 25 GHz is indicated for operating(assuming P=2, Q=3, and IF=2.5G and the first carrier frequency RF=35GHz).

In the single mixing stage example using LO 500-A described above, thefrequency difference between result A and result B is about 1.33 GHzsuch that a VCO with a tuning range as low as about 6.6 about (or about7 percent) with an center operating frequency of around 21 GHz (which isthe frequency midpoint between result A and result B) can be tuned toprovide (for example) a master frequency signal for deriving LO signalsin quadrature for driving mixers used to modulate (or demodulate) andinformation signal.

In various applications, the VCO operating frequency can be selected,and the second mixer stage and the second—e.g.,“downstream”—multiplication-order operation can be programmably bypassedby selectively opening or closing switches (such as switch S₁ of 700-A)under software control or via mask programming. The selection of upperor lower sideband injection is also programmably selected through byselectively opening or closing switches under software control (forexample, operating frequency of the VCO can be altered within the tuningrange of the VCO by switchably changing the L/C value of the VCO timingcircuit).

In a two mixing stage example using LO 500-A, where P=2, Q=3, andIF=2.5G and for a first carrier frequency of 28 GHz, a VCO generating amaster frequency 12.75 GHz is indicated when the in-phase andquadrature-phase mixers are programmed for low-side injection (e.g., ofthe upper sideband). When the in-phase and quadrature-phase mixers areprogrammed for high-side injection (e.g., of the lower sideband), a VCOgenerating a master frequency 15.25 GHz (result X) is indicated foroperating (assuming P=2, Q=3, and IF=2.5G and the first carrierfrequency RF=28 GHz).

In the two mixing stage example using LO 500-A, where P=2, Q=3, andIF=2.5G and for a second carrier frequency of 35 GHz, a VCO generating amaster frequency 16.25 GHz (result Y) is indicated when the in-phase andquadrature-phase mixers are programmed for low-side injection (e.g., ofthe upper sideband). When the in-phase and quadrature-phase mixers areprogrammed for high-side injection (e.g., of the lower sideband), a VCOgenerating a master frequency 18.75 GHz is indicated for operating(assuming P=2, Q=3, and IF=2.5G and the first carrier frequency RF=35GHz).

In the two mixing stage example using LO 500-A described above, thefrequency difference between result X and result Y is about 1 GHz suchthat a VCO with a tuning range as low as about 6.6 about (or about 7percent) with an center operating frequency of around 15.75 GHz (whichis the frequency midpoint between result A and result B) can be tuned toprovide (for example) a master frequency signal for deriving LO signalsin quadrature for driving mixers used to modulate (or demodulate) andinformation signal.

Accordingly, using a combination of one or two mixing stages and withupper and lower sideband (low and high-side injections), a lowerfrequency VCO can be used in the architecture, with moderate tuningrange requirements. Moreover, the center frequency of the VCO is anon-integral submultiple of the RF frequency, which makes thearchitecture resistant to any frequency pulling effects from (e.g.,parasitic coupling by) the PA.

FIG. 6 is a schematic diagram of baseband phase shifter circuits inaccordance with example embodiments. Phase shifters 600-A and 600-B aretwo embodiments of baseband phase shifter elements, which include agenerally cyclical arrangement of resistors (instances of R₁ and R₂) andcapacitor components (instances of C). Phase shifter 600-A is apolyphase phase shifter element for receiving a differential signal(IN+) from the baseband output and for generating quadrature signals(I+, I−, Q+, and Q−) for coupling to the outputs, which are coupled tothe up-conversion mixers (discussed above). Phase shifter 600-B is aconfiguration polyphase phase shifter element for receiving a quadraturedifferential signal (INI+, INI−, INQ+, and INQ−) from the basebandoutput and for generating quadrature signals for coupling to theup-conversion mixers. Target applications include embodimentsimplemented in a system on-chip application, where the baseband signalscan be efficiently generated in quadrature. The polyphase phase shifterelement performs phase shifting, as well as filtering for reducingout-of-band DAC sampling aliasing.

The outputs from the baseband phase shifter element 600-A (or 600-B) arecoupled to trigonometric scaling factor blocks (e.g., 402-A1, 402-A2 to402-N1, 402-N2), which use scaling factors to scale the quadraturebaseband signals. The scalar factors are responsible for the phase shiftprovided by the transmitter element, and are given by tan(φ_(x))=β/α asdiscussed above with reference to Eq. 2. The trigonometrical weights, α,and β, can be implemented as a single attenuator element or a cascade oftwo or more attenuator elements. As an example, cascaded scalarweighting can be implemented as a product of two weight factors α₁, β₁,and α₂, β₂ in a manner that α=α₁α₂, and β=β₁β₂. However, the transmitterelements in a cascaded array can be formed using a same architecture tomaintain similar accuracy and phase balance metrics. A selectedtrigonometric weighting can include unit element-based structures (e.g.,matched unit passive components such as resistors, capacitors,transistors etc.), and a trigonometric number mapped to its closestinteger values (e.g. tan(22.5)=2/5).

FIG. 7 is a schematic diagram of scalar elements for generatingtrigonometric weightage for a vector modulator in accordance withexample embodiments. Passive components (e.g., resistors R andcapacitors C) are arranged in a ladder configuration, such that unitelements are used and tapped at intermediate points (taps A, B, C, D, E,F, G, H, I, and J) to implement the weight factors. For example, ladders700-A, 700-B, and 700-C are three embodiments for implementingtrigonometric weighting. A differential input signal is coupled to(e.g., the left of) the passive chain formed by the ladder and aweighting is selected by closing (e.g., through software control ofprogrammable switches (e.g., S_(1A), S_(1B), S_(2A), S_(2B), S_(3A),S_(3B), S_(4A), S_(4B), S_(5A), and S_(5B)) and/or maskprogramming/rewiring of metal layers of an integrated circuit) a switchassociated with a set of tapping points (e.g. {A,B}, {C,D}, {E,F},{G,H}, and {I,J}, where more or less sets can be used). For example, twopassive networks include similar type of elements (resistors orcapacitors) serially arranged. The ratios α_(k) and β_(k) for the k-th(e.g., a selected) transmitter element are determined by the ratio ofthe impedance presented by the selected tapping point (e.g., B, D, F, H,or J) to that of the total impedance of the passive impedance line(e.g., where the last tapping point is selected):{αk,βk}=Zr/Zt′  (Eq. 14)where Zr is the equivalent impedance at the selected tapping point, andZt is the total impedance of the line. The impedance ratios areapplicable to passive ladder networks including resistive, capacitive,and inductive elements serially arranged as the laddered elements.

FIG. 8 is a schematic diagram of a one-local-oscillator up-conversionmixer in accordance with example embodiments. Positive and negativequadratic baseband input signals for a selected transmitter element arecoupled to a respective gate of first-level transistors 800-A, 800-B,800-C, and 800-D for selectively sinking (e.g., grounding) a respectivecurrent signal. The output currents of first-level transistors 800-A,800-B, 800-C, and 800-D are each dynamically controlled in response tothe phases of respective differential quadrature signals applied to thegates of the second-level transistors (e.g., 801-A, 801-B, 801-C, 801-D,801-E, 801-F, 801-G, and 801-H). The second-level transistors arecontrolled in response to a first local oscillator generating a first LOfrequency in quadrature. The differential output currents generated bycoupling the gates of commonly-gated transistors are combined togenerate to a single-ended beamforming output voltage by adifferential-to-single-ended conversion element (e.g., balun) 802. Theconversion element (e.g., combiner) 802 is for coupling the single-endedbeamforming output voltage to a single-ended power amplifier element 803for transmission via transmitter element antenna 804 (e.g., of a phasedarray of antenna).

Accordingly, a beamforming (e.g., focusing) transmitter element outputsignal is generated in response to a first first-level transistor forsinking a first current signal in response to a positive in-phasebaseband (BB) signal, a second first-level transistor for sinking asecond current signal in response to a negative in-phase BB signal, athird first-level transistor for sinking third current signal inresponse to a positive quadrature-phase BB signal, and a fourthfirst-level transistor for sinking a fourth current signal in responseto a negative quadrature-phase BB signal.

A first second-level transistor is arranged to control a first-directionoutput in response to a first positive in-phase local oscillator (LO)frequency and the first current signal. A second second-level transistoris arranged to control a second-direction output in response to a firstnegative in-phase LO frequency and the first current signal. A thirdsecond-level transistor is arranged to control the first-directionoutput in response to the first negative in-phase LO frequency and thesecond current signal. A fourth second-level transistor is arranged tocontrol the second-direction output in response to the first positivein-phase LO frequency and the second current signal. A fifthsecond-level transistor is arranged to control the first-directionoutput in response to a first positive quadrature-phase LO frequency andthe third current signal. A sixth second-level transistor is arranged tocontrol the second-direction output in response to a first negativequadrature-phase LO frequency and the third current signal. A seventhsecond-level transistor is arranged to control the first-directionoutput in response to the first negative quadrature-phase LO frequencyand the fourth current signal. An eighth second-level transistor isarranged to control the second-direction output in response to the firstpositive quadrature-phase LO frequency and the fourth current signal.

An output combiner is arranged to generate the beam-focusing transmitterelement output signal in response to the first-direction output signaland the second-direction output signal. An amplifier is arranged toamplify the beam-focusing transmitter element output signal fortransmission by a transmitter element antenna. Optionally, the amplifierand the combiner can be combined.

FIG. 9 is a schematic diagram of a one-local-oscillator up-conversionmixer including programmable segmented transistors in accordance withexample embodiments. Positive and negative quadratic baseband inputsignals for a selected transmitter element are coupled to a respectivegate of first-level transistors 900-A, 900-B, 900-C, and 900-D forselectively sinking (e.g., grounding) a respective current signal. Therespective output currents of first-level transistors 900-A, 900-B,900-C, and 900-D are individually and dynamically controlled in responseto the phases of respective differential quadrature signals applied tothe gates of the second-level transistors (e.g., 901-A, 901-B, 901-C,901-D, 901-E, 901-F, 901-G, and 901-H). The second-level transistors arecontrolled in response to a first local oscillator generating a first LOfrequency in positive and negative quadrature. The differential outputcurrents generated by coupling the gates of commonly-gated transistorsare converted to a single-ended voltage by using adifferential-to-single-ended conversion element (e.g., balun) 902, forcoupling to a single-ended power amplifier element 903 for transmissionvia transmitter element antenna 904 (e.g., of a phased array ofantenna).

Each of the second-level transistors (e.g., 901-A, 901-B, 901-C, 901-D,901-E, 901-F, 901-G, and 901-H) is a “segmented” transistor in which“segments” (e.g., paralleled transistors having differing sized channelsand/or transconductances) carry portions of an entire current inparallel. For example, transistor 901-A is shown as a series oftransistor segments 901-A1 through 901-AK, where “K” is the number ofsegments used to form the entire transistor 901-A. Each segment isselected via mask programming (which forms wired connections to selectedtransistor segments 901-A1 through 901-AK) or switches (which formprogrammably selected connections to selected transistor segments 901-A1through 901-AK). In similar fashion, transistors 900 are segmented astransistor segments 900-A through 900-P, where “P” is the number ofsegments used to form the entire transistor 900, and where each segmentis selectively coupled using programmable switches or mask programming.

The trigonometrical weights (α,β) are implemented by selecting varioussegments of each second-level segmented transistor (e.g., 901-A1 through901-AK) and first-level segmented transistor (where each instance ofsegmented transistor 900 includes 900A-900P). Accordingly, particulartrigonometric weights (e.g., for operating at a specific frequency) areprogrammably selected (e.g., such that a unitary system design/chip canbe reused for operating efficiently in various allocated frequency“spectrum”). The balun element 902 converts differential signal to asingle ended signal and the power amplifier element 903 amplifies thesingle ended signal and provides output to the transmitter elementantenna 904. Each of the second-level transistors (e.g., 901-A, 901-B,901-C, 901-D, 901-E, 901-F, 901-G, and 901-H) is independently weighted(e.g., scaled) with trigonometric weights independently of any othersecond-level transistor.

Accordingly, the first current signal (described above with respect toFIG. 8) is generated in response to a first trigonometric weight, thesecond current signal is generated in response to a second trigonometricweight, third current signal is generated in response to a thirdtrigonometric weight, and the fourth current signal is generated inresponse to a fourth trigonometric weight, wherein the first, second,third, and fourth trigonometric weights are individually selectable.

FIG. 10 is a schematic diagram of a two-local-oscillator up-conversionmixer in accordance with example embodiments. Positive and negativequadratic baseband input signals for a selected transmitter element arecoupled to a respective gate of first-level transistors 1000-A, 1000-B,1000-C, and 1000-D for selectively sinking (e.g., grounding) arespective current signal. The respective output currents of first-leveltransistors 1000-A, 1000-B, 1000-C, and 1000-D are individually anddynamically controlled in response to the phases of respectivedifferential quadrature signals applied to the gates of the second-leveltransistors (e.g., 1001-A, 1001-B, 1001-C, 1001-D, 1001-E, 1001-F,1001-G, and 1001-H). The second-level transistors are controlled inresponse to a first local oscillator generating a first LO frequency inpositive and negative quadrature. A third level of transistors includestransistors (e.g., 1002-A, 1002-B, 1002-C, 1002-D, 1002-E, 1002-F,1002-G, and 1002-H) respectively coupled in series with a respectivefirst level transistor and are controlled in response to a second localoscillator generating a second LO frequency in positive and negativequadrature.

The differential output currents generated by coupling the gates ofcommonly-gated transistors are converted to a single-ended voltage byusing a differential-to-single-ended conversion element (e.g., balun)1002, for coupling to a single-ended power amplifier element 1003 fortransmission via transmitter element antenna 1004 (e.g., of a phasedarray of antenna).

Each of the second-level transistors (e.g., 1001-A, 1001-B, 1001-C,1001-D, 1001-E, 1001-F, 1001-G, and 1001-H) is a segmented transistor.For example, transistor 1001-A is shown as a series of transistorsegments 1001-A1 through 1001-AN, where “N” is the number of segmentsused to form the entire transistor 1001-A, where each segment isindividually and selectively coupled using programmable switches or maskprogramming. Likewise, each of the third-level transistors s (e.g.,1002-A, 1002-B, 1002-C, 1002-D, 1002-E, 1002-F, 1002-G, and 1002-H) is asegmented transistor where, for example, transistor 1002-A is shown as aseries of transistor segments 1002-A1 through 1002-AN, where “N” is thenumber of segments used to form the entire transistor 1002-A, andwherein each segment is individually and selectively coupled usingprogrammable switches or mask programming. Similarly transistors 1000are segmented as transistor segments 1000-A through 1000-N, where “N” isthe number of segments used to form the entire transistor 1000, andwhere each segment is selectively coupled using programmable switches ormask programming.

A first set of trigonometrical weights (α,β) are implemented byselecting various segments of each second-level segmented transistor(e.g., 1001-A1 through 1001-AK) and first-level segmented transistor(where each instance of segmented transistor 1000 includes 1000A-1000P).A second set of trigonometrical weights (α,β) are implemented byselecting various segments of each third-level segmented transistor(e.g., 1002-A1 through 1002-AN) and first-level segmented transistor(where each instance of segmented transistor 1000 includes 1000A-1000N).Accordingly, particular trigonometric weights (e.g., for operating at aspecific frequency) are programmably selected (e.g., such that a unitarysystem design/chip can be reused for operating efficiently in variousallocated frequency “spectrum”). The balun element 1002 convertsdifferential signal to a single ended signal and the power amplifierelement 1003 amplifies the single ended signal and provides output tothe transmitter element antenna 1004. Each of the third-leveltransistors (e.g., 1002-A, 1002-B, 1002-C, 1002-D, 1002-E, 1002-F,1002-G, and 1002-H) is independently weighted (e.g., scaled) withtrigonometric weights independently of any other third-level transistor.

Accordingly, a first third-level transistor is arranged to control thefirst-direction output (with reference to the discussion above withrespect to FIG. 8) in response to a second positive in-phase localoscillator (LO) frequency and the first current signal. A secondthird-level transistor is arranged to control the second-directionoutput in response to a second negative in-phase LO frequency and thefirst current signal. A third third-level transistor is arranged tocontrol the first-direction output in response to the second negativein-phase LO frequency and the second current signal. A fourththird-level transistor is arranged to control the second-directionoutput in response to the second positive in-phase LO frequency and thesecond current signal. A fifth third-level transistor is arranged tocontrol the first-direction output in response to a second positivequadrature-phase LO frequency and the third current signal. A sixththird-level transistor is arranged to control the second-directionoutput in response to a second negative quadrature-phase LO frequencyand the third current signal. A seventh third-level transistor isarranged to control the first-direction output in response to the secondnegative quadrature-phase LO frequency and the fourth current signal. Aneighth third-level transistor is arranged to control thesecond-direction output in response to the second positivequadrature-phase LO frequency and the fourth current signal.

FIG. 11 is a schematic diagram of a transmitter elements forintermediate frequency-level trigonometric weighting in accordance withexample embodiments. Each of the baseband signals (e.g., BB₁ to BB_(N))is coupled to a respective polyphase phase shifter network (e.g., PPF1101-A through 1101-N). The baseband-filtered quadrature differentialoutputs from the PPF 1101 are coupled to a first stage 1110 ofquadrature mixers (e.g., 1102-A1, -A2, -A3, -A4 to 1102-N1, -N2, -N3,-N4). The first stage 1110 of quadrature mixers is arranged toup-convert the baseband-filtered quadrature differential signals fromthe PPF 1101. For example, mixer 1102-A1 is arranged to up-convert afiltered in-phase baseband signal in response to an in-phase LO (e.g.,LO_(1I)) signal for generating a first once-modulated in-phase signal.Mixer 1102-A2 is arranged to up-convert a filtered in-phase basebandsignal in response to a quadrature-phase LO (e.g., LO_(1Q)) signal forgenerating a second once-modulated in-phase signal. Mixer 1102-A3 isarranged to up-convert a filtered quadrature-phase baseband signal inresponse to a quadrature-phase LO (e.g., LO_(1Q)) signal for generatinga first once-modulated quadrature-phase signal. Mixer 1102-A4 isarranged to up-convert a filtered quadrature-phase baseband signal inresponse to an in-phase LO (e.g., LO_(1I)) signal for generating asecond once-modulated quadrature-phase signal.

The outputs (the C) from the first stage 1110 of quadrature mixers arecoupled to respective trigonometric weights scaling elements (e.g.,1103-A1, -A2, -A3, -A4 to 1103-N1, -N2, -N3, -N4) to generate post-firststage trigonometrically weighted signals (first and secondonce-modulated in-phase trigonometrically weighted signals, and firstand second once-modulated quadrature-phase trigonometrically weightedsignals). The trigonometric scaling factors at IF frequency (e.g.,1103-A1, -A2, -A3, -A4 to 1103-N1, -N2, -N3, -N4) can be implementedusing serial component(s) such as any combination of 700-A, 700-B and700-C (although serial inductors can be cost-inefficient in manyapplications).

The post-first stage IF trigonometrically weighted signals output by thetrigonometric scaling factor elements are coupled to the single-sidebandcombiner elements (e.g., summation elements 1104-A1, -A2 to 1104-N1,-N2). For example, LO combiner element 1104-A1 performs the operationαI+βQ, whereas LO combiner element 1104-A2 performs the operation αQ−βI.Combiner elements up though 1104-N1 and 1104-N2 perform similaroperations, except using same or different values of α and β for eachtransmitter element. Accordingly, the IF-level αI+βQ and αQ−βI signalsare respectively weighted (in the IF domain by combiners such as 700-A,700-B, and/or 700-C) in accordance with a first set of (IF)trigonometric weights α and β for a particular transmitter element k.Optionally, the trigonometric weights scaling elements (e.g., 1103) andthe combiner elements (e.g., 1104) can be combined.

The outputs (αI+βQ and αQ−βI signals) from the single-sideband combinerelements are provided to the second stage 1120 of mixer elements (e.g.,1105-A1, -A2 to 1105-N1, -N2) to generate twice-LO-modulatedtrigonometrically weighted αI+βQ and αQ−βI signals. Thetwice-LO-modulated once-trigonometrically-weighted signals αI+βQ andαQ−βI output from the second stage 1120 of mixers are coupled to thesecond sideband combiner element (1106-A through 1106-N) to generateRF-modulated (e.g., up-converted) combined αI+βQ+αQ−βI signal. Thecombined αI+βQ+αQ−βI signal is accordingly a pre-amplificationtrigonometrically-weighted signal.

Each of the pre-amplification trigonometrically-weighted signals for aparticular transmitter element k is coupled to a respective poweramplifier (e.g., one of 1107-A through 1107-N). After amplification, theamplified trigonometrically-weighted signals are transmitted (e.g.,through the air) by physically separated antennas 1108-A through 1108-N(e.g., which are separated by a distance including a fraction of awavelength of a transmission carrier frequency).

FIG. 12 is a schematic diagram of receiver elements for basebandtrigonometric weighting in accordance with example embodiments. Receiverarray 1200 includes a number “N” of receiver elements workingconcurrently, where each receiver element receives an RF signal from arespective antenna (1209-A through 1209-N). More specifically, low noiseamplifiers (LNAs 1208-A through 1208-N) are coupled to receive a signal(modulated on carrier wave) from a respective antenna 1209 in an antennaarray.

The output of each LNA 1208 is coupled to a respective first stage 1210of quadrature down-conversion mixers (e.g., 1207-A1, 1207-A2 to 1207-N1,1207-N2). The first stage 1210 of quadrature down-conversion mixers isarranged to down-convert a received RF signal from an LNA 1208: Thefirst stage 1210 of quadrature down-conversion mixers is arranged todown-convert a received RF signal from an LNA 1308: a first mixer1207-A1 is coupled to a first in-phase LO (e.g., LO_(2I)) signal forgenerating a first once-demodulated signal; and a second mixer 1207-A2is coupled to a first quadrature-phase LO (e.g., LO_(2Q)) signal forgenerating a second once-demodulated signal.

The outputs (first and second once-demodulated signals) of the firststage 1210 of quadrature down-conversion mixers (e.g., the first andsecond once-demodulated IF signals) are coupled to the second stage 1220of quadrature down-conversion mixers (e.g., 1206-A1, 1206-A2 to 1206-N1,1206-N2). The second stage 1220 of quadrature down-conversion mixers isarranged to down-convert the first and second once-demodulated signalsignals: a first mixer 1206-A1 is coupled to a second in-phase LO (e.g.,LO_(1I)) signal for generating a first twice-demodulated signal; and asecond mixer 1206-A2 is coupled to a second quadrature-phase LO (e.g.,LO_(IQ)) signal for generating a second twice-demodulated signal.

The outputs (e.g., the first and second twice-demodulated signals) ofthe second stage 1220 of quadrature down-conversion mixers are coupledto a respective trigonometric scaling factor element (e.g., one of1205-A1, -A2, -A3, -A4 to 1205-N1, -N2, -N3, -N4), where eachtrigonometric scaling factor element is a combiner such as combiner700-A or 700-B. For example, a first trigonometric scaling factorelement 1205-A1 is arranged to scale the first twice-demodulated signalin accordance with a selected α (where an α of another receiver elementcan have a different selected value of α) to generate an α-scaledin-phase signal (signal αI), a second trigonometric scaling factorelement 1205-A2 is arranged to scale the first twice-demodulated signalin accordance with a selected β (where a β of another receiver elementcan have a different selected value of β) to generate a β-scaledin-phase signal (signal βI), a third trigonometric scaling factorelement 1205-A3 is arranged to scale the second twice-demodulated signalin accordance with the selected β to generate a β-scaledquadrature-phase signal (signal βQ), and a fourth trigonometric scalingfactor element 1205-A4 is arranged to scale the second twice-demodulatedsignal in accordance with the selected α to generate an α-scaledquadrature-phase signal (signal αQ).

Outputs (signals αI, βI, βQ, and αQ) from the trigonometric scalingfactor elements 1204 are coupled to a single sideband combiner 1204(e.g., 1204-A1, -A2 to 1204-N1, -N2). For example, combiner 1204-A1 isarranged to generate signal αI+βQ in response to signals αI and βQ, andcombiner 1204-A2 is arrange to generate signal αQ−βI in response tosignals αQ and βI.

The outputs (signals αI+βQ and αQ−βI) from the single sideband combiner1204 are coupled to a pair 1203 of low-pass baseband filters (e.g.,1203-A1, -A2 to 1203-N1, -N2). For example, the low-pass baseband filter1203-A1 is arranged to generate a filtered αI+βQ signal in response tothe αI+βQ signal, and the low-pass baseband filter 1203-A2 is arrangedto generate a filtered αQ−βI signal in response to the αQ−βI signal.

The outputs (the filtered αI+βQ signal and the filtered αQ−βI signal)from the pair 1203 of low-pass baseband filters are coupled to a pair1202 of analog to digital converters (e.g., ADCs 1202-A1, -A2 to1202-N1, -N2) for digitizing the received input signals and generating adigital bit stream for each ADC 1202 (for generating in-phase andquadrature-phase digital signals). The digital outputs from all ADCs arecoupled to the digital signal processor (DSP) 1201. Accordingly, each ofthe receiver elements in receiver 1200 uses a (e.g., potentially)different trigonometric scaling factor value between adjacent receiverelements to implement a phase shift of 180/N between signals receivedfrom adjacent antenna structures 1209. Further, each of the bit streamsincludes an output information signal, which includes information fromthe input information signal generated by a respective antenna.

The DSP 1201 is arranged to program and control any programmablecomponent (such as digitally controlled switches) or programmablefunction of the receiver array 1200 and optionally any programmablecomponent/function of a transmitter element, for example when thereceiver array 1200 is part of a transceiver system. For example, theDSP is arranged to control functions such as filtering, phase shifting,trigonometric weighting and beamforming, upper or lower sidebandinjection selection, second mixing stage bypass, multiplier and divisorvalues for the multiplication-level operation unit of the programmablelocal oscillator, controlling transconductance and current density oftransistors through selection of transistor segments in various mixers.

FIG. 13 is a schematic diagram of receiver elements for radio frequencytrigonometric weighting in accordance with example embodiments. Receiverarray 1300 includes a number “N” of receiver elements workingconcurrently, where each receiver element receives an RF signal from arespective antenna (1309-A through 1309-N). More specifically, low noiseamplifiers (LNAs 1308-A through 1308-N) of an element are coupled toreceive a signal (modulated on carrier wave) from a respective antenna1309 in an antenna array.

The output of each LNA 1308 is coupled to a respective trigonometricscaling factor element (e.g., 1307-A1, -A2, -A3, -A4 to 1307-N1, -N2,-N3, -N4), where each trigonometric scaling factor element is a combinersuch as combiner 700-A or 700-B. For example, a first trigonometricscaling factor element 1307-A1 is arranged to scale the received signalin accordance with a selected α (where an a of another receiver elementcan have a different selected value of a) to generate a first α-scaledreceived signal, a second trigonometric scaling factor element 1307-A2is arranged to scale the received signal in accordance with a selected β(where a β of another receiver element can have a different selectedvalue of β) to generate a first β-scaled received signal, a thirdtrigonometric scaling factor element 1307-A3 is arranged to scale thereceived signal in accordance with the selected β to generate a secondβ-scaled received signal, and a fourth trigonometric scaling factorelement 1307-A4 is arranged to scale the received signal in accordancewith the selected α to generate a second α-scaled received signal.

Outputs (the first α-scaled signal, the first β-scaled signal, thesecond β-scaled signal, and the second α-scaled signal) from thetrigonometric scaling factor elements 1306 are coupled to respectivequadrature down-conversion mixers (e.g., 1306-A1, 1306-A2, 1306-A3,1306-A4 to 1306-N1, 1306-N2, 1306-N3, 1306-N4) of the first stage 1310.The first stage 1310 of quadrature down-conversion mixers is arranged todown-convert a received RF signal from an LNA 1308: a first mixer1306-A1 is coupled to a first in-phase LO (e.g., LO_(2I)) signal forgenerating a first once-demodulated trigonometrically weighted signal;and a second mixer 1306-A2 is coupled to a first quadrature-phase LO(e.g., LO_(2Q)) signal for generating a second once-demodulatedtrigonometrically weighted signal.

The outputs (first and second once-demodulated trigonometricallyweighted signals) of the first stage 1310 of quadrature down-conversionmixers are coupled to the second stage 1320 of quadraturedown-conversion mixers (e.g., 1305-A1, 1305-A2 to 1305-N1, 1305-N2). Thesecond stage 1320 of quadrature down-conversion mixers is arranged todown-convert first and second once-demodulated trigonometricallyweighted signal signals: a first mixer 1305-A1 is coupled to a secondin-phase LO (e.g., LO_(1I)) signal for generating a firsttwice-demodulated trigonometrically weighted signal; and a second mixer1305-A2 is coupled to a second quadrature-phase LO (e.g., LO_(1Q))signal for generating a second twice-demodulated trigonometricallyweighted signal.

The outputs (first and second twice-demodulated trigonometricallyweighted signals) of the second stage 1320 of quadrature down-conversionmixers are coupled to a single sideband combiner 1304 (e.g., 1304-A1,-A2 to 1304-N1, -N2). For example, combiner 1304-A1 is arranged togenerate signal αI+βQ in response to signals αI and βQ, and combiner1304-A2 is arrange to generate signal αQ−βI in response to signals αQand βI.

The outputs (signals αI+βQ and αQ−βI) from the single sideband combiner1304 are coupled to a pair 1303 of low-pass baseband filters (e.g.,1303-A1, -A2 to 1303-N1, -N2). For example, the low-pass baseband filter1303-A1 is arranged to generate a filtered αI+βQ signal in response tothe αI+βQ signal, and the low-pass baseband filter 1303-A2 is arrangedto generate a filtered αQ−βI signal in response to the αQ−βI signal.

The outputs (the filtered αI+βQ signal and the filtered αQ−βI signal)from the pair 1303 of low-pass baseband filters are coupled to a pair1302 of analog to digital converters (e.g., ADCs 1302-A1, -A2 to1302-N1, -N2) for digitizing the received input signals and generating adigital bit stream for each ADC 1302 (for generating in-phase andquadrature-phase digital signals). The digital outputs from all ADCs arecoupled to the digital signal processor (DSP) 1301. Accordingly, each ofthe receiver elements in receiver 1300 uses (e.g., slightly) differenttrigonometric scaling factor values between adjacent receiver elementsto phase shift signals received from adjacent antenna structures 1309,where the adjacently received (e.g., by adjacent receiver elements)signals by 180/N.

FIG. 14 is a flow diagram for selection of values for first and secondmultiplication-order operations for a local oscillator in accordancewith example embodiments. In a first embodiment program flow 1400-A, forexample, values of P and Q are selected in accordance with two step upmixing used for transmitting (or step down is used for receiving) wheretandem multiplication-order operation units are arranged in adivide-first-then-multiply configuration using a first even orderdivider (/P) including and output coupled to a second odd-ordermultiplier (XQ).

In operation 1401, two center frequencies of RF carriers are selected asRF₁ and RF₂, with the initial constraint of RF₁<RF₂. For example, whenRF₁=28 GHz and RF₂=35 GHz, the two center frequencies of 28 GHz and 35GHz are accordingly targeted as RF carrier frequencies.

In operation 1402, an initial value for each of P and Q are determined.For example, the values of each of P and Q can be determined byselecting and/or adjusting each value over a relatively limited numberof iterations of operations 1402, 1403, and 1404.

In operation 1403, corresponding VCO frequencies VCO₁ and VCO₂ and atuning range TR are determined. For example, VCO frequencies VCO₁ andVCO₂ are determined in accordance with the equations

${VCO}_{1} = {\frac{P}{Q + 1}{RF}\; 1}$and

${VCO}_{2} = {\frac{P}{Q - 1}{RF}\; 2}$and the effective tuning range is determined in accordance with theequation

${TR} = {\frac{Q + 1}{Q - 1}{\left( \frac{{RF}\; 2}{{RF}\; 1} \right).}}$

In operation 1404, a decision is made as to whether the tuning range TRis within an acceptable tuning range (TR_(ACC)). For example, theacceptable tuning range is the tuning range over which a selected VCOoperates within parameters selected for a particular application. Whenthe current values for P and Q do not acceptably fall within the tuningrange, program flow proceeds to operation 1402, where a (e.g., further)adjustment in P and Q is made such that individual values of P and Qsatisfying the acceptable tuning range are converged upon. When thecurrent values for P and Q acceptably fall within the tuning range,program flow proceeds to terminus 1405, where the determined values of Pand Q are retained and subsequently used to program a local oscillator(such as 500-A and 500-B).

In a second embodiment program flow 1400-B, similar steps to programflow 1400-A are executed. For example, values of Q and P are selected inaccordance with two step up mixing used for transmitting (or step downis used for receiving) where tandem multiplication-order operation unitsare arranged in a multiply-first-then-divide configuration using a firstodd-order multiplier (XQ) including an output coupled to a second evenorder divider (/P).

In operation 1406, two center frequencies of RF carriers are selected asRF₁ and RF₂, with the initial constraint of RF₁<RF₂. For example, whenRF₁=28 GHz and RF₂=35 GHz, the two center frequencies of 28 GHz and 35GHz are accordingly targeted as RF carrier frequencies.

In operation 1407, an initial value for each of P and Q are determined.For example, the values of each of P and Q can be determined byselecting and/or adjusting each value over a relatively limited numberof iterations of operations 1407, 1408, and 1409.

In operation 1408, corresponding VCO frequencies VCO₁ and VCO₂ and atuning range TR are determined. For example, VCO frequencies VCO₁ andVCO₂ are determined in accordance with the equations

$\left( {{VCO}_{1} = {\frac{P}{Q\left( {P + 1} \right)}{RF}\; 1}} \right.$and

${{VCO}_{2} = {\frac{P}{Q\left( {P - 1} \right)}{RF}\; 2}},$the effective tuning range is determined in accordance with the equationusing

${TR} = {\frac{P + 1}{P - 1}{\left( \frac{{RF}\; 2}{{RF}\; 1} \right).}}$

In operation 1409, a decision is made as to whether the tuning range TRis within an acceptable tuning range (TR_(ACC)). For example, theacceptable tuning range is the tuning range over which a selected VCOoperates within parameters selected for a particular application. Whenthe current values for P and Q do not acceptably fall within the tuningrange, program flow proceeds to operation 1402, where a (e.g., further)adjustment in Q and P is made such that individual values of Q and Psatisfying the acceptable tuning range are converged upon. When thecurrent values for Q and P acceptably fall within the tuning range,program flow proceeds to terminus 1405, where the determined values of Qand P are retained and subsequently used to program a local oscillator(such as 500-A and 500-B).

What is claimed is:
 1. A circuit, comprising: a multiplication-orderoperation unit for receiving a master frequency signal, for generating afirst derived frequency signal in response to the master frequencysignal in accordance with a first multiplication-order operation, andfor generating a second derived frequency signal in response to thefirst derived frequency signal in accordance with a secondmultiplication-order operation, wherein the second multiplication-orderoperation is a multiplication-order operation that is inverse to thefirst multiplication-order operation; a first mixer stage for mixing inquadrature an input information signal in response to the first derivedfrequency signal to generate a first mixer stage output signal includingcomponents of the first derived frequency signal; a trigonometricweighting scaler for trigonometrically scaling the input informationsignal in response to a trigonometric weight to generate a scaler outputsignal including components of the trigonometric weight; a second mixerstage for mixing in quadrature the input information signal in responseto the second derived frequency signal to generate a second mixer stageoutput signal including components of the second derived frequencysignal; and an output combiner for generating an output informationsignal, wherein the output information signal includes components of thefirst mixer stage output signal, the scaler output signal, and thesecond mixer stage output signal.
 2. The circuit of claim 1, wherein theinput information signal is received via either an antenna or adigital-to-analog converter (DAC).
 3. The circuit of claim 1, whereinthe second mixer stage output signal is generated in response to thefirst mixer stage output signal.
 4. The circuit of claim 1, wherein thefirst and second mixer stage are arranged to mix in quadrature inaccordance with either a modulation operation or a demodulationoperation.
 5. The circuit of claim 1, wherein the first and second mixerstage are arranged to mix in quadrature in accordance with:cos(ω_(LO1)+ω_(LO2)+ω_(BB))t=cos(ω_(LO1)t)cos(ω_(LO2)t)cos(ω_(BB)t)−sin(ω_(LO1)t)sin(ω_(LO2)t)cos(ω_(BB)t)−sin(ω_(LO1)t)cos(ω_(LO2)t)sin(ω_(BB)t)−cos(ω_(LO1)t)sin(ω_(LO2)t)sin(ω_(BB)t),where ti is time, ω_(LO1) is the first derived frequency signal inquadrature input signal having a frequency higher than a basebandfrequency, ω_(LO2) is the second derived frequency signal in quadratureinput signal and ω_(BB) is the information signal in quadrature at thebaseband frequency.
 6. The circuit of claim 1, wherein the outputcombiner is arranged to combine components of the first mixer stageoutput signal and the second mixer stage output signal in accordancewith a current mode of operation.
 7. The circuit of claim 1, comprisinga sideband combiner arranged to generate a single sideband outputinformation signal in response to the second mixer stage output signal.8. The circuit of claim 1, wherein the trigonometric weighting scaler isarranged to selectively phase shift the output information signal. 9.The circuit of claim 1, wherein the first multiplication-order operationis executed by converting energy from fundamental frequency to odd-orderharmonics and wherein the second multiplication-order operation isexecuted by one or more divide-by-two digital dividers.
 10. The circuitof claim 1, wherein the first multiplication-order operation and thesecond multiplication-order operation are programmable.
 11. The circuitof claim 1, comprising a voltage controlled oscillator (VCO) forgenerating the master frequency signal operable at a frequency lowerthan a carrier frequency of the output information signal generated bythe output combiner.
 12. The circuit of claim 11, wherein the VCO centeroperating frequency is a sub-harmonic of the carrier frequency of theoutput information signal generated by the output combiner.
 13. Thecircuit of claim 1, comprising a sideband combiner arranged to generatea single sideband output information signal in response to high-sideinjection of lower sidebands of the developed sidebands when a lowerfrequency carrier frequency is selected, and to generate the singlesideband output information signal in response to low-side injection ofhigher sidebands of the developed sidebands when the higher frequencycarrier frequency is selected, wherein the output combiner generates theoutput information signal in response to the single sideband outputinformation signal, and wherein the selected carrier frequency is afrequency of either the input information signal or the outputinformation signal.
 14. A system, comprising: a voltage controlledoscillator (VCO) for generating a master frequency output signal inaccordance with a selection of either a lower frequency carrierfrequency or a higher frequency carrier frequency, wherein the masterfrequency output signal is generated at a first VCO frequency when thelower frequency carrier frequency is selected, wherein the masterfrequency output signal is generated at a second VCO frequency higherthan the first VCO frequency when the higher frequency carrier frequencyis selected, and wherein the VCO includes a tuning range encompassingthe first and second VCO frequencies; a multiplication-order operationunit for generating local oscillator signals in quadrature in responseto the maser frequency output signal; and a first one or more mixerstages for developing sidebands in response to a first receivedinformation signal and the local oscillator signals in quadrature,wherein the one or more mixer stages are arranged to generate an outputinformation signal in response to high-side injection of lower sidebandsof the developed sidebands when the lower frequency carrier frequency isselected, and wherein the first one or more mixer stages are arranged togenerate the first output information signal in response to low-sideinjection of higher sidebands of the developed sidebands when the higherfrequency carrier frequency is selected.
 15. The system of claim 14,comprising: a second one or more mixer stages for developing sidebandsin response to a second received information signal and the localoscillator signals in quadrature, wherein the one or more mixer stagesare arranged to generate a second output information signal in responseto high-side injection of lower sidebands of the developed sidebandswhen the lower frequency carrier frequency is selected, and wherein thesecond one or more mixer stages are arranged to generate the secondoutput information signal in response to low-side injection of highersidebands of the developed sidebands when the lower frequency carrierfrequency is selected.
 16. The system of claim 15, wherein the first oneor more mixer stages are arranged to phase shift components of the firstoutput information signal with respect to a first antenna in an antennaarray, and wherein the second one or more mixer stages are arranged tophase shift components of the second output information signal withrespect to a second antenna in the antenna array.
 17. The system ofclaim 15, wherein the first one or more mixer stages are arranged tophase shift components of the first input information signal withrespect to a first antenna in an antenna array, wherein the second oneor more mixer stages are arranged to phase shift components of thesecond input information signal with respect to a second antenna in theantenna array, wherein the first input information signal is receivedfrom the first antenna in the antenna array, and wherein the secondinput information signal is received from the second antenna in theantenna array.
 18. The system of claim 17, including a processor forprocessing the first output information signal in accordance with thephase shifted components of the first input information signal, forprocessing the second output information signal in accordance with thephase shifted components of the second input information signal, toarrange the first and second one or more mixers for the high-sideinjection of the lower sidebands of the developed sidebands when thelower frequency carrier frequency is selected, and to arrange the firstand second one or more mixers for the low-side injection of the highersidebands of the developed sidebands when the higher frequency carrierfrequency is selected.
 19. A method, comprising: generating a masterfrequency output signal in accordance with a selection of either a lowerfrequency carrier frequency or a higher frequency carrier frequency,wherein the master frequency output signal is generated at a firstvoltage controlled oscillator (VCO) frequency when the lower frequencycarrier frequency is selected, wherein the master frequency outputsignal is generated at a second VCO frequency higher than the first VCOfrequency when the higher frequency carrier frequency is selected, andwherein the VCO includes a tuning range encompassing the first andsecond VCO frequencies; generating local oscillator signals inquadrature in response to the maser frequency output signal; anddeveloping sidebands in a first one or more mixer stages in response toa first received information signal and the local oscillator signals inquadrature, wherein the one or more mixer stages are arranged togenerate an output information signal in response to high-side injectionof lower sidebands of the developed sidebands when the lower frequencycarrier frequency is selected, and wherein the first one or more mixerstages are arranged to generate the first output information signal inresponse to low-side injection of higher sidebands of the developedsidebands when the higher frequency carrier frequency is selected. 20.The method of claim 19, comprising: developing sidebands in a second oneor more mixer stages in response to a second received information signaland the local oscillator signals in quadrature, wherein the one or moremixer stages are arranged to generate a second output information signalin response to high-side injection of lower sidebands of the developedsidebands when the lower frequency carrier frequency is selected, andwherein the second one or more mixer stages are arranged to generate thesecond output information signal in response to low-side injection ofhigher sidebands of the developed sidebands when the lower frequencycarrier frequency is selected.